We've created the following video to help in the understanding of this topic:

There are no plans right now for a 75 Ω VNA. However, the PicoVNA 106 and software do support 75 Ω measurement, either:

- mathematically using nominal Z
_{0}impedance conversion - using a port impedance adaptor and a 75 Ω calibration kit

If you are working with 75 Ω systems you may already have a suitable port impedance adaptor and cal kit.

Time-domain network analysis and frequency domain network analysis are very similar measurements. The former applies a spectrum of discrete frequencies to the unknown network: a step or impulse incident waveform is applied and an oscilloscope or sampling head captures the reflected and transmitted waveforms. The latter applies a series of discrete frequencies and captures reflected and transmitted amplitude and phase using phase-sensitive (IQ) receivers.

The frequency-domain VNA approach has better dynamic range because the applied power at each frequency can be constant and relatively high, and the receivers can have restricted noise bandwidth.

TDR/TDT can theoretically be quicker because a single step or impulse could give all necessary information. However, the high sampling-time resolution required by this method tends to call for a sequential sampling oscilloscope such as the PicoScope 9311. This captures only one sample point on each cycle of a step or impulse, so only repeating signals can be tested. Even so, our TDR or TDT solution is still slightly quicker than the VNA. See below regarding multiple forward, reverse, transmission, and reflection measurements.

The PicoVNA vector network analyzer and the Pico 9311 TDR/TDT Sampling Oscilloscope solutions are comparable on primary features and performance:

- In the context of cable or transmission line testing, time-domain measurements have the advantage of directly showing impedance continuity v time (TDR) or actual pulse response (TDT). Assuming propagation velocity is reasonably well known, a direct interpretation or readout of impedance v distance is possible. Fortunately, the PicoVNA includes a time-domain readout without extra cost, so both solutions can achieve the ideal readout.

- Where it is necessary to combine measurements with other system elements (measured or simulated) then scattering parameters, Smith charts, etc. tend to be preferred. While S-parameters and time domain measurements can be interrelated by an FFT, only the PicoVNA supports both output formats.

- The VNA is limited to 6/8.5 GHz and best effective time resolution of 120/85 ps. Assuming a line propagation velocity of 2c/3, this will resolve impedance v distance along your line to around 24/17 mm [with short/open fault location around five times better). The PicoScope 9311 fast step has a system transition time of around 60 ps, giving it roughly the same time (2 x 60 ps) and distance resolution, giving it roughly the same resolution as the PicoVNA 106.

- At its best time domain resolution, the PicoVNA can test at 4096 k discrete sample points. This translates to a maximum path length of around 100 m. At lower resolution, longer path lengths can be accommodated. The longest path length supported by the PicoScope 9311 is theoretically limited by the widest available pulse width of 4 µs, which translates to 400 m in a reflection measurement. In practice, however, at this length, the resolution of a TDR measurement will be lost to high-frequency cable losses and a VNA may match or even outperform it. TDR distance can be increased by instead using the lower frequency capability of PG900 fast step generators.

The larger differentiators between the two are:

- The VNA automatically sequences reflection and transmission measurements and can do this in both directions through an unknown 2-port network. Greater speed and much-reduced manual intervention are significant benefits when more than a single port reflection or a single transmission measurement is required. Single-direction measurement is typically sufficient for transmission line measurements, but for devices with transmission losses or gain, bidirectional measurements are far more commonly needed and the automated VNA wins even more convincingly.

- In differential line applications the PicoScope 9311 is likely to win. It has differential de-skewable step sources and its two channels can be used to receive either differential reflection or differential transmission. In the case of a differential line without a nearby ground, such as a twisted pair in free space, the PicoScope 9311 can determine differential impedance v distance from a single measurement setup. The differential test stimulus develops a virtual ground (or signal null) at the midpoint between the conductors. The VNA can only stimulate one conductor with respect to the other and cannot resolve differential impedance. It could, however, determine an impedance for a given core and how that changes with length, and could be used in comparison with a gold standard—for instance, both conductors separately measured, the other grounded— in this case.

If we now consider the opposite extreme example of a differential line: the individually screened twinax pair. Here a ground surrounds and isolates the two lines of the differential pair. In this case, the two differential cores can each be measured separately as individual coaxial lines. Their differential impedance will be the sum of the individual impedances. The PicoScope 9311 and the VNA can both do this in a single measurement setup. The two ports of the VNA can be used to separately measure the two cores, but only one TD readout is permitted at a time.

Of course, a great many differential lines lie somewhere between these two extremes with varying degrees of ground and pair coupling. The usefulness and accuracy of the VNA measurement vary accordingly.

Pico is uniquely positioned to support either solution and will be happy to support your decision-making for a given application.

Not at present, but we would be interested to know your requirements.

The SMA and PC3.5 connectors on the PicoVNA make ideal relatively low-uncertainty ports from which to port-adapt both to the larger format and legacy interfaces and to the emerging smaller format and point-contact interfaces. In many cases, port adaptors can achieve sufficiently low measurement uncertainty for systems with connector types of poorer repeatability, and in other applications where performance is not limited by the connector.

Where measurement demands are high, you may be able to obtain port adaptors with either de-embed data or reference plane offset values; both these correction mechanisms being supported by the PicoVNA. In the absence of data, you can purchase a mating pair of port adapters such as PC3.5(f/m) to N(m) and PC3.5(f/m) to N(f). Mate the pair at the N-ports and measure them, then attribute half of the transmission parameters and an average of the reflection parameters to each. This is often how the manufacturer measures adaptors that are provided with data.

Ideally, the adapted ports would be calibrated with port-matching cal kits. These are available from a range of suppliers.

Likewise, within-series and adapting test leads are even more widely available than cal kits. However, it seems likely that emerging small-format connectors will demand a lighter-gauge, more flexible test lead or a flexible port adaptor. Please let us know if you have special requirements in this area.

Yes - but only the PicoVNA 108, not the PicoVNA 106.

In most circumstances, a vector network analyzer stimulates a device port in signal frequency steps and measures all the ports at the same frequency steps. However, there are applications where you need to measure at harmonics of the stimulus. There are also applications, around frequency mixer devices, where you need to measure at a frequency (the intermediate frequency, or IF) that is offset from the stimulus.

Using the example of the PicoVNA 106, we have:

- 140 kHz bandwidth
- 10 MHz to 6 GHz sweep
- 201 points (12-term cal): 37 ms for full s2p result set. = 184 µs per point for forward AND reverse sweep. > 5400 points/sec
- 201 points (S21 cal): 25 ms for single S21 result set. = 124 µs per point for a forward sweep. > 8000 points/sec

Be careful when comparing with competitors, as they tend only to quote figures for the single-point single sweep. That compares with our faster figure above and not with the slightly longer time for a forward and reverse sweep necessary for a full set of S-parameters. This product is amongst the faster units available. See the competitive comparison on the web site under the **Reviews** tab.

The sweep speed reduces according to bandwidth settings as the measurement at the output of bandwidth filter has to settle to its full accuracy.

If we first set a 10,001 pt sweep, the forward/reverse switch is proportionally less of an interruption and we can estimate its duration from the two trace lengths that we have.

At 140 kHz 170 us / pt for full s2p (two sweeps):

- Forward / reverse switch implied at around 14 µs.

Testing at other bandwidths for 10k point sweeps:

- At 10 kHz: 430 µs/pt for full s2p (two sweeps)
- At 1 kHz: 3 ms/pt
- At 100 Hz: 28.5 ms/pt
- At 10 Hz: 285 ms/pt

Think of a modern communications protocol and the amplitude/phase constellation that it uses to represent data. For instance, here's a QAM constellation:

Each of these data codes is transmitted and received on an RF signal or carrier with an amplitude and a phase.

When we send such a signal through any amplifier, but particularly a transmitting power amplifier, distortions will lead to problems.

If the amplifier is operated towards its peak power capability, non-linearity, usually compression, will cause the long vectors in the diagram (above) to end up shorter than they should be, the shorter vectors less affected. The corners will move inwards. If the degree of compression is –1 dB compared with a fully linear amplifier, our vector length will be down around 10%.

This is a single point on a curve, but it helps to understand the whole curve. Our P1dB utility plots this curve at a given frequency by sweeping the input power to the device:

If at the same point, or any other amplitude, the phase of the signal shifts, those constellation points will be affected. Generally again, the high-power ones will move around the circumference of a circle. When this phase shift is amplitude- (vector length) related we have *phase modulation due to amplitude modulation* or *AM to PM conversion*. The outer constellation points move around a circle, the inner ones less so and the constellation becomes twisted.

Clearly these amplifier distortions are important, and can be critical in any modulated RF system more advanced than Morse Code - even FM to a degree. If a data code is too far misplaced, it cannot be recognized and corruption occurs. Once an amplifier or any potentially nonlinear element has been characterized for these distortions, it will either be used within its limitations or, much more commonly now, the input will be predistorted to compensate the device. Why? To save battery life! We need to drive low-power amplifiers as far into compression as we possibly can. It is hard to imagine just how much work has been done in this area!

One further point. A great many comms signals hop around the carrier frequency, and the distortions in a device vary with frequency. That is why our P1dB utility can perform the measurement at up to 201 different frequencies in a sweep of frequency and power.

Our recommendation is 1 Nm for stainless steel connectors of type N, SMA or PC3.5 as supplied by us. We offer two torque wrench accessories with our recommended setting:

- The TA356 SMA wrench will fit all SMA, PC3.5 and K-type connectors, which are standardized at 8 mm across flats.
- The TA358 N wrench will fit and operate reliably on all 20 mm and most 19 mm nuts (some 19 mm are too rounded at the corners and will need a 19 mm wrench). Our wrench will not fit the majority of 13/16 inch (20.6 mm) nuts.

If mating to gold-plated brass, you should use a reduced torque of around 0.56 Nm to avoid damage to the softer connector. A wrench with this torque setting is readily available from other suppliers.

- Preset torque limit: 1 N·m (8.85 in·lb)
- Across flats: 8 mm

- Preset torque limit: 1 N·m (8.85 in·lb)
- Across flats: 20 mm

TRL (through-reflect-line) calibration is a complex subject. TRL calibration and de-embedding are two separate mechanisms for the removal of fixture errors within a measurement. In fact there are at least five mechanisms that may be appropriate, depending upon the fixture, its errors, the measurement and the accuracy required:

**Reference plane shift or rotation**– Removes the effect of delay in the fixture feed lines, but not their loss, mismatch loss or flatness errors. Applies for transmission (S21, S12) and reflection (S11, S22) measurements.**Normalization**– Removes the effects of loss and flatness of the feedlines for transmission measurements.**Time-domain gating**– Assuming that time-domain translation is available for the accurate*time*location of fixture errors, an additional time-domain gating function can be used to gate them out and then convert back to the frequency or s-parameter domain. This is effective for reflection measurements and removes fixture mismatch and delay, but not fixture loss.**De-embedding**– Uses S-parameter data files for each of the test port feedlines. These are mathematically backed out of (de-embedded from) all transmission and reflection measurements, wholly removing their errors (at least to the level of accuracy in those de-embed files).**TRL calibration**– While we can readily and accurately define and apply SOL (short-open-load) calibration standards to a connectorized test port, we can't so readily define SOL standards to apply at the end face of any given fixture feedline. If we could, we would be able to define a calibration reference plane right at the DUT terminals and remove all errors to the feed side of that reference plane. TRL calibration instead uses lengths of transmission line that have identical structure to the feed lines, often several lengths (to achieve a broadband calibration) and with various terminations; a practical, if inconveniently time-consuming, process and one that cannot fully address fixture mismatch errors.

The PicoVNA supports mechanisms (1), (2), (1)+(2) and (4).

Despite having the time-domain function available, we do not have the gating function that would support (3). Nor do we support (5) in its many complex forms that in practice become necessary.

Our User's Guide has some good background and of course instruction in the use of reference plane shift, normalization and de-embedding. This will help you to decide which method is most appropriate to your application.

If you have to apply de-embedding to your measurement, there are two options for the derivation of the de-embed s-parameter files. Most commonly a simulation model of the feedlines would be used. You might also have a measurement option. If the feedline and launch points are geometrically identical, for instance, you might measure Port 1 and 2 feeds as a through pair and attribute half of the transmission parameters and an average of the mismatch parameters to each. The choice depends on the details of your fixture, your available fixture test pieces and your application.

Pico specifies its noise floor with a substantial margin. The specification is –100 dB to 10 MHz, –110 dB to 4000 MHz and then –100 dB to 6000 MHz.

The specification limits apply to a noise quantity known as "displayed average noise". The user in need of a deep noise floor will however need to know more about the actual noise level likely to be seen and perhaps what the numbers mean. That is what we shall discuss here.

All VNA manufacturers quote, in addition to a specification limit, a typical noise floor level. However, different definitions of noise are seen and there can be a lack of clarity in meaning.

So, first of all, let us take a "typical" PicoVNA 106 and either of the Pico test leads. We calibrate an S21 measurement at 10 Hz bandwidth (Isolation and Through) using either of the Pico calibration standards. We then terminate the two ports and sweep S21 from 300 kHz to 6 GHz in 1001 points, and we will see something like this:

We have used 1001 points to ensure that we see sufficient spectral detail; to do so is disadvantageous in a comparison with the common practice of using 201 points. Statistically this will reveal higher noise peaks. Meaningful comparison is the intent here, not seeming to be the best.

Pulling that noise trace into Excel (blue) and then adding 15 more sweeps at exactly the same settings (yellow) we see noise statistics at work as more and more data points are added. Noise peaks are sitting at around –104 dB with 16 000 data points in the picture opposite:

Let the blue trace join the 15 yellow traces and then do some maths. The plot opposite shows:

- RMS noise level (dark blue). Here, the log scale values are converted to a linear attenuation factor and then the root mean square of the 16 noise traces is taken and converted back to the log scale of the plot. This trace is a measure of noise power that is present and will be important to those wanting to make noise measurements on the VNA.
- Average noise level (light blue). Again we convert to linear attenuation factor, average the 16 traces and then convert back to log. This is the true average level of noise present in the measurement. Often quoted, but not necessarily stated to have been calculated accurately.
- Displayed average noise (purple). To be honest, this is not a meaningful measure at all; it has simply become common practice to quote and specify it (not least because it gives lower numbers). It is a linear average of the 16 traces as they appear on the display – irrespective of the logarithmic scale that they are plotted against. There is validity here in that most VNAs do include a noise averaging feature that works in this way to give a quieter trace.

We have added two specification lines and a guide line:

- Solid red – PicoVNA 106 guaranteed specification – displayed average noise
- Dashed red – PicoVNA 106 typical specification – displayed average noise
- Dashed black – typical level for noise power in the measurement – RMS noise

Median and average difference between RMS and average noise is 1.0 dB in these plots. Max 2.1 dB.

Median and average difference between RMS and displayed average noise is 2.3 dB. Max 4.7 dB.

Finally, the relatively high specification margin applied by Pico takes into account that the PicoVNA 106 is an all-new design for which statistical multiple batch data is being compiled. While no unit will be worse than the limit, there is no implication that a given unit is likely to be close to the limit. The typical performance really is typical of what you should expect from the product.

*When I try to calibrate to 50 Ω the Smith chart says 51 Ω and when I do an open calibration the impedance is off the Smith chart. Is there a problem with my PicoVNA or the calibration file?*

The calibration standards are not perfect and do not need to be.

To begin with, they have finite length – so reflections from the short and open must take finite time to travel from the reference plane to the short/open and back to the reference plane. In our case the standards are physically long and the effective length is about 17 mm (SMA standards), or a bit shorter for PC3.5 standards . If in the **Enhancements Menu** you add Reference Plane shift of about 16 to 17 mm, the Smith chart will display short and open more closely at the short/open points. The explanation is that 17 mm is about 85 ps travel time given a wave velocity of about 5 ps / mm. There and back is therefore 170 ps. At 6 GHz that is about 360 degrees on the Smith chart, so our short and open display as a near full-unity circle on a full-span frequency sweep. After reference plane shift is added, the standards still do not display as a perfect short, open and load. This is because they are still not perfect zero, infinite or matched impedance – they all have finite imperfections – and this is most clearly revealed at higher frequencies.

Calibration works, therefore, through accurate knowledge of the standards' characteristics when tested on a reference VNA that is directly traceable to national standards. Then their length and their impedance imperfections are accurately captured. That accurate knowledge is captured in the .kit file that is supplied with the calibration kit. That file is loaded to the VNA to tell the calibration process exactly what the short, open and load should measure. You can inspect a .kit file with a text editor and will find it to be a concatenation of the s-parameters for each of the four SOLT elements. Full details are given in the NMT Kit User Guide Appendix 2.

The data that we supply with our kits is better than that supplied with most available calibration standards. Typically, full data, unique to serial number, is supplied by others only with premium kits. Competitors tend to use instead only polynomial models, or sometimes only average models on lower-cost kits. We instead supply full and unique data with all our Premium and Standard kits and this allows more accurate calibration.

In particular, note that when polynomial models are used, these do not model the load. This places great need upon the load standard to be very good (and therefore expensive) and for the VNA to assume that it is perfect (having nothing to define otherwise) and to display it as a perfect dot on the Smith chart. This is of course is impossible at any price and is in fact an erroneous display.

So perfect dots on a Smith chart should almost never be expected or trusted. A a perfect dot implies significantly less than 1% loss or reflection and for the short, open and through significantly less than 1° phase delay at the maximum frequency of the sweep. To do that, you would need something that sits wholly inside the test port connector right at its reference plane and to have no physical size of its own!

Above are typical plots of our Premium Female PC3.5 Calibration Standards out to 8.5 GHz.